Data communication method and apparatus

ABSTRACT

A data stream is impressed on a carrier by forming from the data stream a succession of binary symbol words which control the modulation level of the carrier during respective modulation time intervals. To generate the symbol words from the data stream, the latter is first divided into a base bit stream (14) and one or more second bit streams (15). These second streams (15) are then subjected to error control coding before being used to define the least significant bits of the symbol words; the most significant bits are provided by the base bit stream. As a result, although the least significant bits of the transmitted symbol words (IV,III,II,I) are more prone to noise corruption than the most significant bits this tendency is compensated for by the error coding employed. The overall effect is to minimize power requirements while retaining good bandwidth efficiency.

The present invention relates to a data communication method andapparatus and in particular, but not exclusively, to a data modem foruse in a satelite communication system.

BACKGROUND OF THE INVENTION

With the increasing use of satellites for business data communicationsthere is a growing need for data modems with good efficiency ofbandwidth and power.

Recent developments in high efficiency modems have tended to concentrateon power efficiency. Thus a modern data modem may take bits one or twoat a time from a data stream that has been previously subject to ForwardError Control, and use these bits to control the level of phasemodulation of a carrier during a corresponding modulation time interval,the resultant transmitted symbols each having two or four possiblelevels (this modulation process is generally referred to as two-orfour-level phase shift keying, PSK). These modern modems have broughtdown the required satellite power by a factor in excess of 4 to 1 (6dB)with respect to the requirements of only a few years ago. However, thisimproved power efficiency has been at the expense of occupied bandwidth,which, allowing for adjacent channel guardbands, can be up to 3 or 4times the data rate. At lower data rates there is sufficient bandwidthavailable for this to be accommodated but in the 2 or 4 Mbit/sec datarate range, bandwidth efficiency becomes much more critical.

One method of improving bandwidth efficiency is to increase the numberof possible levels of each symbol to increase the number of transmittedbits/Hz of occupied bandwidth. However as the number of levels persymbol is increased, the symbol power has to be increased dramaticallyto prevent noise from increasing the bit error rate. The need for thisdramatic increase is due to the fact that as the numbers of levels persymbol is increased, the different symbol combinations are notorthogonal to each other as in the two or four level symbol case andthere is degradation similar in effect to cross-talk.

It is an object of the present invention to provide a data communicationmethod and apparatus with good bandwidth and power efficiency.

SUMMARY OF THE INVENTION

According to one aspect of the present invention, there is provided amethod of data communication including transmitting a binary data streamby a process including the steps of:

dividing the data stream into a first, base bit stream and at least onesecond bit stream;

subjecting the said at least one second bit stream to error controlcoding to produce at least one overlay bit stream, and

utilising said base and overlay bit streams to generate a succession ofsymbol words of binary form for controlling the level of modulation of acarrier signal during respective modulation time intervals, the mostsignificant bit or bits of each symbol word being derived from the basebit stream while the least significant bit or bits are derived from thesaid at least one overlay bit stream.

It will be appreciated that the least significant bits of the symbolwords are those most likely to be corrupted by noise during theirtransmission as modulation of the carrier (that is, as a modulationsymbol) since the margin for error is least for these bits. In thepresent system, the fact that these least significant bits have beensubjected to error control coding enables their successful recovery at areceiver at much lower signal/noise ratios than would otherwise bepossible. Thus an increase in bits/transmitted symbol can be achievedwithout the same power level increase penalty as would be incurred withstandard multi-level symbol systems.

Further according to the invention, there is provided a method of datacommunication including receiving data transmitted in accordance withthe last but one paragraph, the receiving process including the stepsof:

generating a succession of modulation-level binary words representingthe level of modulation of the carrier signal during respectivemodulation time intervals,

deriving from the least significant bits of the modulation-level wordsat least one received overlay bit stream corresponding to the said atleast one overlay bit stream generated during the transmission process,

effecting error correction on the said at least one received overlay bitstream in accordance with the error control coding implemented in thetransmission process,

removing the bits of the said at least one overlay bit stream, aftererror correction, from the modulation-level words and then utilising thelatter to derive a received base bit stream corresponding to the saidbase bit stream generated during the transmission process,

utilising the said at least one error-corrected received overlay bitstream to generate at least one received second bit stream correspondingto said at least one second bit streams generated during thetransmission process, and

combining the received base bit stream and the at least one receivedsecond bit stream to form a received data stream.

Where during the transmission process, two or more second bit streamsare divided from the data stream, the security of the error controlcoding to which each stream is subjected is preferably made the greater,the less the significance of the system word bits determined by thestream concerned; the purpose of this is to compensate for thecorrespondingly higher risk of corruption of the least significant bitsdue to noise. In the receiving process, the received overlay bit streammade up of the least significant bit or bits will generally begenerated, error corrected and removed from the modulation-level wordsbefore the second received overlay bit stream is detected and processed;this similarly applies to the other received overlay bit streams wherepresent.

During the transmission process, the base bit stream may itself besubject to error control coding. Furthermore, the or each overlay bitstream may be divided into further streams at least one of which issubject to further error control coding before being used to determinethe value of bits of the symbol word.

Generally, the bits of the base bit stream will be used one or two at atime to determine the most significant bit or bits of each symbol word.The bits of the or each overlay bit stream will generally be used onebit at a time to determine a corresponding bit of the symbol word.

The error control coding employed may be of the block or convolutionaltype. Preferably, however, an orthogonal or bi-orthogonal block code isused. Furthermore, the symbol words are preferably used to control phaseshift keying of the carrier signal.

Advantageously, the binary data stream is divided into said first base,stream and two said second streams, the second streams being error codedusing respective codes whereby to produce respective said overlay bitstreams which are then used one bit a time in generation of saidsuccession of symbol words, the codeword length of said codes used inproducing each overlay stream being the greater the lesser thesignificant in each symbol word of the bit contributed by that overlaystream.

In one embodiment of the invention, a data stream to be transmitted istaken eighty-six bits at a time, sixty four bits being used to form thebase bit stream, sixteen bits to form one said second bit stream and sixfor another second bit stream. The sixteen bits of the said one secondbit stream are converted, in groups of four, using a bi-orthogonal codeof length eight to form a first overlay bit stream of thirty two bits.The six bits of the said another second bit stream are converted, usinga bi-orthogonal code of length thirty two, into a second overlay bitstream also of thirty two bits. Thirty-two 4-bit symbol words are thenformed by taking two bits from the base stream for the two mostsignificant bits of each word, one bit from the first overlay stream toform the third most significant bit, and one bit from the second overlaystream to form the least significant bit of the symbol word. During thereceiving process, the second overlay is first detected, subjected toerror correction by correlation with all possible bi-orthogonal codewords of length thirty two, and then subtracted from themodulation-level binary words: next the first overlay is detected, errorcorrected and subtracted, leaving binary words representing four-levelsymbols determined by the bits of the base bit stream. The errorcorrected overlays are decoded and recombined with the base stream.

In another embodiment, three second bit streams are formed and are errorcoded using bi-orthogonal codes respectively of length four, sixteen andsixty four. As with the preceding embodiment, the base stream is takentwo bits at a time, and the overlay streams one bit at a time, to formthe symbol words.

According to another aspect of the present invention, there is provideddata communication apparatus for transmitting a binary data stream,characterised in that said apparatus comprises:

data-stream dividing means for dividing the data stream into a first,base bit stream and at least one second bit stream,

error-coding means arranged to subject the said at least one second bitstream to error control coding to produce at least one overlay bitstream, and

a signal organiser arranged to receive said base and overlay streams andto generate therefrom a succession of symbol words of binary form forcontrolling the level of modulation of a carrier signal duringrespective modulation time intervals, the most significant bit or bitsof each symbol word being divided from the base bit stream while theleast significant bit or bits are derived from the said at least oneoverlay bit stream.

According to a further aspect of the present invention, there isprovided data communication apparatus arranged to receive carrier-signalbase data transmitted in accordance with the method set out in the lastpreceding paragraph but ten, said apparatus comprising:

demodulator means for generating a succession of modulation-level binarywords representing the level of modulation of the carrier signal duringrespective modulation time intervals.

overlay detector means for deriving from the least significant bits ofthe said modulation-level words at least one received overlay bit streamcorresponding to the said at least one overlay bit stream generatedduring the transmission process, the overlay detector means beingarranged to effect error correction on the said at least one receivedoverlay bit stream in accordance with the error control codingimplemented in the transmission process,

overlay subtractor means for removing the bits of the said at least oneoverlay bit stream, after error correction, from the modulation-levelwords,

base-stream decoder means arranged to utilise said modification-levelwords, after subtraction of said overlay stream bits, to derive areceived base bit stream corresponding to the said base bit stream

generated during the transmission process,

overlay decoder means arranged to utilise the said at least oneerror-corrected received overlay bit stream to generate at least onereceived second bit stream corresponding to said at least one second bitstream generated during the transmission process, and combining meanscombining the received base bit stream and the at least one receivedsecond bit stream to form a received data stream.

BRIEF DESCRIPTION OF THE DRAWINGS

Several forms of data communication modem embodying the presentinvention will now be described by way of example, with reference to theaccompanying diagrammatic drawings, in which:

FIGS. 1A and 1B are diagrams illustrating the possible phase statesrespectively of a four-level and eight-level phase shift keyed (PSK)carrier;

FIG. 2 is a schematic diagram of the transmitter section of a prior artmodem employing four-level PSK;

FIGS. 3 and 4 are schematic diagrams of the transmitter and receivingsections respectively of a first form of modem embodying the presentinvention;

FIG. 5 is a Table showing the encoding of three data bits asbi-orthogonal codewords of length 4 and also illustrating thecross-correlation characteristics of these codewords;

FIG. 6 is a phase state diagram illustrating the detection of the leastsignificant bit of a transmitted symbol;

FIG. 7 is a Table illustrating the correlation of stored and receivedcodewords effected in the receiving section illustrated in FIG. 4;

FIGS. 8 and 9 are schematic diagrams of the transmitter and receivingsections respectively of a second form of modem embodying the invention;

FIG. 10 is a Table showing the encoding of five date bits asbi-orthogonal codewords of length 16;

FIG. 11 is a block diagram of a phase modulator suitable for use in thetransmitter section of the first and second forms of modem;

FIG. 12 is a diagram illustrating the effect of carrying outinterpolation between phase states in the FIG. 11 phase modulator; and

FIG. 13 is a block diagram of a phase demodulator suitable for use inthe receiving section of the first and second forms of modem.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Before describing the two data modems embodying the present inventionthat are shown in FIGS. 3, 4 and 8, 9 a brief description will be given,with reference to FIGS. 1A and 2, of the transmitter section of a priorart data modem.

The FIG. 2 modem utilises four-level phase shift keying (sometimesreferred to as quadrature phase shift keying or QPSK). A data bit stream10 to be transmitted is fed to a signal organiser block 11 which formsbinary symbol words therefrom by taking bits two at a time (as indicatedby the number in square brackets in block 11) from the bit stream. Thesymbol words may each be considered as representing the value of anangle φ; since each symbol word is composd of two bits, there are fourpossible values for the angle φ. The symbol words are fed in turn to aphase modulator 12 where they determine the phase modulation of acarrier signal of frequency f_(c) during respective modulation timeintervals, the modulator 12 outputting a series of modulation symbolseach having one of four possible phase states depending on the value ofφ. During each modulation time interval, the modulator output can beexpressed as:

    A cos (2πf.sub.c t+φ+K)

where A and K are constants. The arrangement of the modulator 12 isgenerally such that the four possible values of φ are spaced by 90° asindicated in FIG. 1A for the case where K=45°.

By way of example, typical values for three symbol words are indicatedin FIG. 2 together with the resultant transmitted modulation symbols.

The decoding of the modulation symbols in the receiving section (notillustrated) of a receiving modem can be simply effected by deriving abinary word representing the phase modulation value (φ+K) and thendetermining the nearest one of the four possible phase states. Where thebinary words representing the modulation level have a full rangecorresponding to (φ+K) varying from 0° to 360°, then with K=45°, thedecoding simply reduces to utilising the first two bits of each binaryword as these bits represent the angle quadrant of the received symbol.

An eight-level PSK modem takes a form very similar to that of the QPSKmodem of FIG. 2 except that each binary symbol word is formed by threebits from the bit stream to be transmitted, and each transmitted symbolhas eight possible phase states, typically distributed as illustrated inFIG. 1B. Due to the fact that in eight-level PSK the angular spacingbetween possible phase states is half that of QPSK, if the same noiseimmunity is to be achieved then the signal power must be higher foreight level PSK than for QPSK.

The embodiment of the invention shown in FIGS. 3 and 4 will now beconsidered. FIG. 3 illustrates in schematic form the transmittingsection of a data modem. The transmitting section includes a phasemodulator 13 arranged to receive a succession of three-bit symbol words,representing an angle φ, and to output, during respective modulationtime intervals, corresponding modulation symbols having one of eightpossible phase values (φ+K) where K is a constant generally equal to 45°(see FIG. 1B); the phase modulator 13 thus acts as a standardeight-level PSK modulator. It is in the generation of the symbol wordsthat the FIG. 3 modem differs markedly from a standard eight-level PSKmodem.

As is diagrammatically depicted in FIG. 3, the data bit stream to betransmitted is divided in block 8 into bit streams, namely a first, orbase, bit stream 14 and a second bit stream 15. For each eleven bits ofthe data stream, eight bits are used to form the base stream 14 andthree bits are used for the second stream 15. In FIG. 3, the base bitsare shown as the first eight bits of an eleven bit `frame` of the datastream; this positioning of the base-stream bits within the frame isnot, however, critical and bits from any position within the frame canbe used to make up the eight base-stream bits.

The base stream 14 is passed to one input of a signal organiser 16 whichtakes two bits at a time from this stream (as indicated by the figure insquare brackets) to form the two most significant bits (MSBs) ofsuccessive symbol words.

The second bit stream 15 is subjected to error control coding beforebeing passed as an overlay bit stream 17 to a second input of the signalorganiser 16, the bits of this overlay stream being taken one at a timeto form the least significant bit of each successive symbol word.

In the present embodiment, the error control coding to which thesecond-stream bits are subjected, involves the conversion of each groupof three second-stream bits into a corresponding biorthogonal codewordof length four. The two leftmost columns of the FIG. 5 Table show thecorrespondence between the possible values of three data bits and thebiorthogonal codewords.

A property of biorthogonal codewords is that when any codeword iscorrelated with all codewords of the same length on the basis that a bitmatch is valued as (+1) and a mismatch as (-1), then a correlation peakequal in value to the codeword length is produced when the codeword iscorrelated with itself whereas all other correlations sum to zero-withthe exception of the correlation with the inverse codeword where anegative correlation peak is produced. This property is illustrated inthe righthand columns of the FIG. 5 Table where the codewords `1001` and`0101` are both correlated against the whose set of codewords of lengthfour (note in this Table the bit correlation values of (+1) and (-1)have been abbreviated to `+` and `-`). The correlation properties of thebiorthogonal codewords is used in the receiving section of the modem andwill be considered in more detail hereinafter.

The three second-stream bits derived from each elevenbit data-streamframe are thus converted in an overlay coder block 18 to a four-bitbiorthogonal codeword which is output as the overlay bit stream 17.

The signal organiser 16 operating in the manner already described, takesthe eight base-stream bits derived from one data-stream frame and thecorresponding four overlay-stream bits, to form four three-bit symbolwords, the least significant bits of which are determined by the bits ofthe overlay bit stream. The four symbol words in turn control thegeneration of four modulation symbols I, II, III, IV.

By way of example, arbitary values have been assigned to the bits of thedata-stream frame illustrated in FIG. 3; the resulting values of thebase and overlay streams, of the symbol words and of the transmittedsymbols are also shown.

FIG. 4 schematically illustrates the receiving section of the datamodem. This receiving section includes a demodulator 20 arranged todetect the angle of phase modulation of successively received symbols I,II, III, IV of the modulated carrier, the demodulator outputting amulti-bit (for example 8-bit) binary word indicative of the modulationangle (φ+K). The resolution of the demodulator output words is greaterthan that of the symbol words.

The first operation carried out on the demodulator output words is thedetermination therefrom of the state of the LSB of each correspondingtransmitter symbol word whereby to reconstitute the overlay bit stream(albeit possibly with noise induced errors).

The detection of the LSB state of each symbol word from thecorresponding demodulator output word is effected in block 21 in themanner described below with reference to FIG. 6. Considering the casewhere the two MSBs of a symbol word are `00`, the two possible values ofthe symbol word are then `000` and `001` or, in terms of modulationangle (φ+K), +45° and +90°. By comparing the actual received angle(φ+K), as represented by the binary word output from the demodulator,with a reference angle, also represented by a binary word, lying midwaybetween the values +45° and +90° (that is, an angle of 771/2° -see lineR1 in FIG. 6), it is possible to place a value on the likelihood of theLSB of the symbol word being a `1` or a `0`. In perfect conditions, anLSB of `1` would result in a difference angle between the reference R1and the actual received phase angle of +221/2° whereas an LSB of `0`would give a difference angle of -221/2°. In fact, the presence of noiseis likely to make the detected difference angle different from thosevalues.

The assessment of the LSB state for the other combinations (01, 10, 11)of MSBs of a transmitter symbol word is carried out in a similar mannerrelative to angle references R2, R3, R4 (see FIG. 5). Of course, at thestage of LSB detection in the receiving section of the data modem, thevalues of the two MSBs of the symbol word are unknown and therefore eachreference R1, R2, R3, R4 must be subtracted in turn from the modulationoutput word, the smallest difference angle being taken as the oneindicating the LSB state of the corresponding symbol word. The LSBassessment process can, for example, be effected using a microprocessoror, alternatively, by means of a ROM decoder addressed by thedemodulator output and programmed with the LSB state corresponding toeach possible input.

Once the LSB of each of the four symbol words corresponding to an 11-bitframe of the transmitted data stream has been detected, the overlaycodeword constituted by these four LSB's is subject to error correctionby correlation with each of the eight possible codewords of length four,these words being held in a store 22.

As already noted, this correlation should produce a positive correlationpeak when the received overlay codeword matches the stored codeword withwhich it is being correlated, all other correlations summing (ideally)to zero except for that between the received codeword and the inversestored codeword where a negative correlation peak is produced.

The Table shown in FIG. 7 illustrates the correlation process both forideal and typical reception of the overlay codeword `0101` embedded inthe transmission illustrated in FIG. 3. Under ideal conditions, thedifference angle between the modulation angle (φ+K) of the first symbolI (this angle being that produced by the first symbol word `101`, thatis an angle of 270°) and the corresponding reference R3, is +221/2°;similarly for the symbols II, III and IV the difference angles are-221/2°, +221/2°, -221/2° respectively.

In the correlation process, each stored word is taken in turn andcorrelated with the received overlay codeword as represented by thedifference angles (represented in binary form). Where a bit of thestored codeword being correlated has a value `1`, the difference anglerepresenting the corresponding bit of the received overlay codeword ismultiplied by `+1` whereas when the stored codeword bit value is `0`,the corresponding difference angle is multiplied by `-1`. Once thedifference angles have been weighed in this manner, they are summed toproduce the correlation sum for the codeword under consideration. TheFIG. 7 Table shows this correlation process for two stored words only,namely "1001" and "0101"; as can be seen from the ideal case shown inthe left-hand half of the FIG. 7 Table, where the received overlaycodeword and stored codeword correspond (codeword "0101") a correlationsum of 90° is produced whereas for the out-of-correspondence case(codeword "1001") a zero sum is produced.

(It should be noted that in the heading of the FIG. 7 Table thecodewords are set down in the same manner as in FIG. 3, that is with thefirst transmitted bit in the right hand-most position; for this reason,the symbols have been set down in reverse, with the first to betransmitted being lowermost).

Of course, the purpose of the correlating process is to facilitatecodeword detection in the presence of noise, and the right-hand half ofthe FIG. 7 Table illustrates the correlation process in the case wherethe received modulation angles (φ+K) of the four symbols I, II, III, IVdiffer from their transmitted values to the extent indicated by thereferenced crosses in FIG. 6. It is clear from the FIG. 7 Table thateven in the presence of significant noise, the correlation process stillpermits the correct detection of the overlay codeword.

After detection, with error correction, of the codewords constitutingthe overlay bit stream, this overlay is removed from the binary wordsoutput by the demodulator to effectively transform these words from onesrepresenting an eight-level phase modulation into words representing afour-level phase modulation. This is achieved in block 23 by subtractingfrom the demodulator output word a binary value corresponding to 45°when the corresponding bit of the overlay stream is a binary "1" whileleaving the demodulator output word unchanged when the correspondingoverlay stream bit has a value "0". Conceptually, this corresponds toreversing the rotation imposed by the LSB of each transmitter symbolword on the four-level symbol that represents the two MSB of the symbolword.

The binary words output from the overlay subtractor block 23 eachrepresent the two MSB's of a corresponding symbol word formed in thetransmitting section, these bits having been derived from the base bitstream. As has already been described with reference to the prior artQPSK system of FIG. 2, it is a relatively simple matter to determinewhich bit pairing (00,01,10,11) is represented by each word output fromblock 23 and to reconstitute the base bit stream; this process iseffected in block 24 of FIG. 4.

The error-corrected codewords of the overlay bit stream output from theblock 21, as well as being fed to the overlay subtractor block 23, arealso fed to an overlay decoder block 25 where each four-bit codeword istranslated into the corresponding three data bits. The outputs of thebase detector block 24 and overlay decoder block 25 are then recombinedas illustrated in block 9 of FIG. 4 to reconstitute successive frames ofthe original data stream.

It will, of course be appreciated that the various blocks of thereceiving section shown in FIG. 4 primarily represent functions in thereceiving process rather than actual physical hardware blocks (though,of course, a receiver having a physical block form similar to FIG. 4could be constructed). In fact, since the functions of the blocks 21 to25 are all concerned with processing binary words, these blocks can beconveniently implemented by means of a microprocessor as will beapparent to persons skilled in the relevant art.

From the foregoing description of the data modem shown in FIGS. 3 and 4,it can be seen that eleven data bits have been transmitted in fourmodulation symbols that is, 2.75 bits/symbol. While each transmittedsymbol has eight possible states, the noise immunity of the system ismuch better than with standard eight-level PSK because the most noisevulnerable bits of information, (that is, those represented by the LSBof each transmitted symbol word) have been subjected to error controlcoding prior to transmission which greatly increases their noiseimmunity.

The correct operation of the modem receiving section shown in FIG. 4,and in particular the proper operation of the overlay detector 21,depends on the receiving section being able to identify the start ofeach new overlay codeword carried by the incoming symbols. One way ofidentifying to the receiving section that a new codeword is about tobegin would be to insert a brief pause between the transmission of thesymbols associated with each eleven-bit frame of the data stream. Ratherthan inserting a pause between each frame, a more economic arrangementwould be to insert a pause after every N frames. Suitable circuitryprovided in the receiving section could then be used to identify thispause and synchronise the operation of the receiving section with theincoming stream of symbols. The design of suitable detection circuitryis well within the scope of persons skilled in the art and willtherefore not be considered further herein.

An alternative way of arranging for the receiving section to identifythe start of each overlay codeword is to correlate each stored codewordwith successive groupings of four overlay bits, each successive groupinglosing one bit at one end and gaining one at the opposite end ascompared with its predecessor. One in four groupings will correspond toa codeword and the correlation of this grouping with its correspondingstored codeword will produce a high correlation sum. By detecting thegrouping for which such a good correlation is achieved, it is possibleto determine the boundaries of the received overlay codewords.

The second embodiment of the invention will now be described withreference to FIGS. 8 and 9 of the accompanying drawings. This secondembodiment is very similar to the first embodiment already describedexcept that a second overlay bit stream is additionally generated andgaining one at the opposite end as compared with its predecessor. One infour groupings will correspond to a codeword and the correlation of thisgrouping with its corresponding stored codeword will produce a highcorrelation sum. By detecting the grouping for which such a goodcorrelation is achieved, it is possible to determine the boundaries ofthe received overlay codewords.

The second embodiment of the invention will now be described withreference to FIGS. 8 and 9 of the accompanying drawings. This secondembodiment is similar to the first embodiment already described but herea second overlay bit stream is additionally generated and used todetermine a fourth bit of each symbol word in the transmitter section ofthe modem.

As is illustrated in FIG. 8, the data stream to be transmitted isprocessed in 49-bit frames. Thirty-two bits of each frame are dividedoff in block 8 to form a base bit stream 40; twelve bits are used toform one second bit stream 41; and five bits are used to form anothersecond bit stream 42. For each frame, the twelve bits of thefirst-mentioned second bit stream are taken three at a time and coded inblock 43 as orthogonal codewords of length four to form a first overlaybit stream 44 in the same manner as for the overlay bit stream 17 of theFIG. 3 modem. The five bits of the other second bit stream 42 are codedin block 45 as a single bi-orthogonal codeword of length sixteen to forma second overlay bit stream 46.

The base bit stream 40, the first overlay bit stream 44, and the secondoverlay bit stream 46 are fed to respective inputs of a signal organiser47. This organiser 47 forms for each forty-nine bit frame of the datatream, sixteen four-bit symbol words each made up of two bits from thebase stream 40, one bit from the first overlay bit stream 44, and onebit from the second lverlay bit stream 46. The two bits from the basebit stream 40 constitute the two most significant bits (MSB) of thecorresponding symbol word while the single bit taken from the first andsecond overlay streams 44, 46 respectively constitute the third MSB andthe least significant bit (LSB) of the symbol word. The generated symbolwords are fed to a phase modulator 48 where they determine the phasestate of respective modulation symbols, each symbol having sixteenpossible states as determined by the four bits of the correspondingsymbol word.

Of course, with sixteen possible states for each transmitted symbol,there is much greater chance of noise induced errors being present atthe receiving modem for the least significant bit of the symbol wordbeing communicated by each symbol. It is in order to overcome thisincreased susceptibility to noise of each LSB transmitted, that thebi-orthogonal code words used in coding these LSB's are of a muchgreater length than used for the second least significant bits; in otherwords, the second overlay bit stream uses a bi-orthogonal code of a muchgreater length than used for the first overlay bit stream in order tooffset the increases susceptibility to noise experienced by the bits ofthe second overlay stream during transmission. It can be shown thatadequate compensation for the increase of susceptibility to noise can beachieved by using for the second overlay stream bi-orthogonal codewordsfour times the length of those used for the first overlay stream. It isfor this reason that bi-orthogonal codewords of length sixteen are usedin the present example to form the second overlay bit stream 46.

The Table shown in FIG. 10 lists the thirty two possible bi-orthogonalcodewords of length sixteen together with the five data bits representedby each codeword.

The receiving section of the second embodiment is shown in FIG. 9 andincludes a phase demodulator 20 outputting binary words indicative ofthe phase modulation angle (φ+K) of each successively received symbol.The receiving section operates by detecting and subtracting the secondand first overlay bit streams from the demodulator output words beforeconverting the latter into the base bit stream. The operation of theFIG. 9 receiving section is thus analogous to the operation of the FIG.4 receiving section except that the second overlay bit stream is firstdetected and subtracted from the demodulator output words. Thisdetection of the second overlay bit stream is effected in block 50, thisblock 50 operating in substantially the same manner as the block 21 inFIG. 4. Similarly, the block 50 also effects error correction by acorrelation process involving correlating each codeword group of sixteenbits of the detected second overlay bit stream with the set of storedbi-orthogonal codewords of length sixteen, this process being analogousto the process carried out by the block 21 of FIG. 4, the onlydifference being the length of the codewords used.

The error-corrected second overlay bit stream 51 output from the block50 is subtracted in subtractor 52 from the demodulator output wordswhereafter these words are processed in identical manner as for the FIG.4 receiving section. For this reason, the further processing of thewords output by the subtractor 52 that form the base bit stream andfirst overlay bit stream will not be described, reference being directedto the corresponding description given in relation to the FIG. 4receiving section; furthermore, the functional blocks effecting thefurther processing in the FIG. 9 receiving section have been given thesame references as used for their equivalents in FIG. 4.

The codewords of the second overlay bit stream 51, as well as being fedto the subtract block 52, are also fed to a second ovelay decoder 53which converts these codewords into the corresponding data bits inaccordance with the FIG. 10 Table.

The bit streams output by the base detector 24, and the two overlaydecoders 25 and 53 are then combined in block 9 to reconstitute theoriginal data stream.

In the second embodiment, the determination of the codeword boundariesin the received overlay bit streams can be achieved by the same generalmethods employed for the first embodiment. Where a correlation techniqueis used, this is, of course, applied to the overlay stream first to beisolated so that in the present example, successive groupings of sixteenbits from the second overlay stream are correlated in turn with eachstored codeword.

The modem of FIGS. 8, 9 differs from that of FIGS. 3, 4 by theutilisation of an additional overlay bit stream. It is in fact possibleto provide for the generation and utilisation of one or more furtheroverlay bit streams additional to the first and second overlay streamsof the modem of FIGS. 8, 9. Thus, for example, a third overlay bitstream may be generated for the FIG. 8 transmitter section by takinggroups of seven data bits and converting them into correspondingbi-orthogonal codewords of length sixty four (the code word length beingfour times that used to generate the second overlay bit stream forreasons already discussed). With this latter arrangement, a data streamframe of two hundred and three bits is converted into sixty fourfive-bit symbol words with the two most significant bits of each wordbeing taken from a base bit stream constituted by one hundred and twentyeight of the data-stream bits, the third most significant bit of eachword being taken from the first overlay bit stream constituted bysixteen bi-orthogonal codewords of length four derived from forty eightdata-stream bits, the fourth most significant bit of each word beingtaken from a second overlay stream constituted by four bi-orthogonalcodewords of length sixteen derived from twenty data-stream bits, andthe least significant bit of each word being taken from a third overlaybit stream constituted by one bi-orthogonal codeword of length sixtyfour derived from seven data-stream bits. In the receiving section ofthe modem, the third overlay bit stream is removed first followed by thesecond and first overlay streams. Where a fourth and possibly furtheroverlay bit streams are used, then these are also removed in turn in thereceiving section starting with the overlay stream corresponding to theleast significant bit of the transmitted symbol words.

It is not, of course, essential that the first overlay bit stream bederived using bi-orthogonal codes of length four. Thus, for example, ina variant of the modem of FIGS. 8, 9 the first overlay bit stream isgenerated from bi-orthogonal codewords of length eight while the secondoverlay stream is generated from bi-orthogonal codewords of lengththirty two. In this latter arrangement, the data stream bits are takenin frames of eighty-six bits, sixty-four bits being used to form thebase bit stream, sixteen bits being converted in groups of four intofour codewords of length eight constituting the first overlay stream,and six bits being converted to a codeword of length thirty two formingthe second overlay stream. The bits of the base, first overlay andsecond overlay streams are then combined to form thirty-two four-bitsymbol words.

Indeed, the error coding process performed on the or each second bitstream divided from the data stream need not be by way of bi-orthogonalcodes as in the two embodiments described with reference to FIGS. 3, 4and 8, 9. Provided the required level of noise protection is provided itis in fact possible to use any suitable form of error control codingincluding both block and convolutional codes.

It is also to be noted that while in the embodiments described, bits aretaken in pairs from the base stream and singly from the overlay streamto form the symbol words, it is possible to vary this arrangement withany appropriate number of bits being taken from each stream to form eachsymbol; it is envisaged, for example, that in certain applications onlyone bit will be taken from the base stream as well as from the overlaystreams in forming each control word. Where two or more bits of anoverlay stream are used for each symbol word then the security of theerror control coding used for that overlay stream should beappropriately high as the spacing between possible phase states is beingreduced by a proportionally larger amount.

Other variations are possible to the embodiments of the inventionillustrated in FIGS. 3, 4 and 8, 9. Thus in arrangements where two ormore overlay bit streams are utilised, each overlay stream other thanthe first may be derived by dividing out bits from the preceding overlaystream, rather than direct from the data stream, and subjecting thesedivided out bits to a further error control coding process.

It will be appreciated that the data stream itself may be subjected toerror control coding before division into first and second streams.Furthermore, the base stream may also be subject to error controlcoding. Finally, while in the illustrated embodiments the symbol wordsare used to control phase modulation of a carrier it is, of course,possible to use the words to control other types of carrier modulationsuch as frequency modulation.

Having discussed in detail the formation of the symbol words inaccordance with the invention and their detection in the modem receivingsection, a description will now be given of a suitable form of phasemodulator and demodulator for the modems of FIGS. 3, 4 and 8, 9. FIG. 11shows a phase modulator arranged to receive the binary-form symbolwords. As already mentioned, these words can be considered asrepresenting an angle φ and serve to control the phase modulation of acarrier of frequency f_(c) by an angle (φ+K). To this end, the symbolwords are fed in turn to a read only member 60 arranged to output, foreach input symbol word, first and second binary words, on buses 61 and62, which respectively represent the values of the functions cos (φ+K)and sin (φ+K) for the symbol word concerned.

The outputs of the ROM 60 are thus two digital streams each having aword rate equal to the symbol rate. In order to effect spectrum shaping,the ROM outputs are fed to a pair of digital filters 63, 64 which may beof any suitable form. In the present example, the filters 63, 64 areconstituted by interpolation filters the output clock rates of which area multiple of the symbol rate. The symbol rate, the output clock rate ofthe interpolation filters, and the digital filter spectral response areselected for a given data rate, by a data rate controller 65.

The output of each digital filter 63, 64 is fed to a respectivedigital/analogue converter 66, 67 and from there via a respective lowpass filter 68, 69 to one input of a respective mixer 70, 71.

A local oscillator 72 is arranged to generate a signal A cos 2πf_(c) twhere the value of f_(c) is typically 70 MHz. This signal is fed directto a second input of the mixer 70, that is, the mixer receiving thefiltered signal cos (φ+K). The output of the mixer 70 is thus cos(2πf_(c) t).cos(φ+K).

The output of the local oscillator 72 is also fed, via a 90° phaseshifter 73, to a second input of the mixer 71 the first input of whichis arranged to receive the filtered signal sin (φ+K). The output of themixer 71 is thus -sin (2πf_(c) t).sin(φ+K).

The outputs of the two mixers 70 and 71 are added together in a summer74 the output of which is passed through a bandpass filter 75 toeliminate out-of-band spurious signals and harmonics that may have beengenerated by the mixers. The output of the filter 75 is the signal acos(2πf_(c) t+φ+K).

Generally the output of the filter 75 constitutes the output of the IFstage of the modem transmitter section.

Where the FIG. 11 phase modulator forms part of a modem intended for usewith a communications satellite, then additional components arepreferably provided to make the signal output therefrom better suited tothe characteristics of the transponder of the communications satellite.Satellite transponders are generally built around travelling wave tubepower amplifiers; due to the non-linear characteristic of such anamplifier it is best to only transmit constant envelope signals throughthe transponder. However, a signal that has constant envelope at alltimes usually has a very wide bandwidth and it is therefore generallynecessary to restrict the signal bandwidth in order to preventinterference with adjacent channels. With reference to the FIG. 11 phasemodulator, a side effect of the two bandwidths limiting filters 63, 64is to introduce envelope variations which are exacerbated by certainsequences of phase states (for example, by successive symbols being inanti-phase relation).

One way of dealing with this problem is to interpolate the sequence ofphase shifts to restrict the bandwidth of the phase-modulator outputsignal without introducing envelope variations. This can be achieved byfronting the ROM 60 with an interpolator (not shown) receiving symbolwords at a rate, say, of 1/T and outputting interpolated words at amultiple m/T of the symbol rate. The interpolator interpolates in anon-linear fashion between phase states represented by the symbol words.This process is illustrated in FIG. 12 where the phase states ofsuccessive symbol words are represented by solid lines and thecorresponding interpolated phase values are shown in dashed lines.

Another way of avoiding adverse envelope variations is to control thegeneration of symbol words in such a manner that certain sequences ofphase states are forbidden. Such an arrangement would be particularlysimple to implement where a microprocessor is used to process the datastream to be transmitted, the microprocessor functioning to generate thebase and overlay bit streams and to organise them into symbol words. Byarranging for the microprocessor to compare successive symbol words witheach other it is possible to prevent the output of symbol word sequencesproducing undesirable phase state transmissions. It should be noted thatthe prevention of certain phase sequences does in fact serve to reducethe data capacity of the overall system.

FIG. 13 shows one possible form of the phase demodulator for the modemreceiving sections shown in FIGS. 4 and 9. In addition to the phasedemodulator components, FIG. 13 also shows as block 80 the base andoverlay stream detector-decoder circuitry of the receiving section, anda multiplexer block 81 for reconstituting the transmitted data streamfrom the base and overlay streams.

The signal fed to the phase demodulator will generally be an IF signal.This signal is fed to the input of two mixers 82, 83 to which are alsofed sin and cos signals at the carrier signal frequency f_(c), these sinand cos signals being produced by a voltage controlled oscillator 84 anda phase shifter 85. The output of the oscillator 84 is kept in phaselock with the incoming signal by means of a phase lock loop. This loopoperates on a signal produced by subtracting the detected base streamfrom the signal fed to the base detector 24 (see FIGS. 4 and 9); thesignal produced by this subtraction is a measure of the phase errorpresent due to noise plus the phase error due to imperfect tracking ofthe received signal by the oscillator 84. In FIG. 13, the generation ofthis phase error signal by the above-mentioned subtraction process iseffected in block 86. By averaging out the noise phase error in a loopfilter 88, after having first converted the phase error signal into ananalogue voltage via a digital to analogue converter 87, it is possibleto derive a signal indicative of phase lock error which can be fed backto the oscillator 84 to reduce this error. As an alternative to theforegoing coherent detection arrangement, it is, of course, possible touse differential PSK to encode the symbol words onto the carrier; inthis case, the actual received value of (φ+K) would correspond to thedifference between successive output values of ROM 94 in FIG. 13. Wheredifferential PSK is used, the blocks 86, 87 and 88 are no longerrequired.

The output of the two mixers 82, 83 are fed via respective analogue todigital converters 89, 90 to respective digital filters 91, 92 thecharacteristics of which are controlled by a data rate controller 93 tomatch the characteristics of these filters to those of the filters 63,64 of the phase modulator. The outputs of the filters 91 and 92respectively represent, in binary form, the values sin(φ+K) and cos(φ+K). These binary output signals are fed to respective inputs of aread only memory (ROM) 94 arranged to output a binary signal on bus 95equal in value to the quantity (φ+K). The binary signal on bus 95 isthen fed to block 80 for processing in the manner already described.

From the foregoing, it is apparent that modems embodying the inventiongive improved data rate/bandwidth efficiency characteristics as comparedwith standard PSK systems for the same power levels and error rates.Thus, for example, the described embodiment using two overlays codedwith bi-orthogonal codeword of lengths eight and thirty-tworespectively, has a data rate/bandwidth efficiency of;

    2+(4/8)+6/32=2.6875 bits per Hz

as compared with two bits/Hz for an equivalent standard QPSK system. Theembodiment of the invention using three overlays with codeword lengthsof four, sixteen and sixty-four gives a data rate/bandwidth efficiencyof 3.17 bits/Hz.

We claim:
 1. A method of data communication including transmitting abinary data stream by a process characterised by the steps of:dividingthe data stream into a first, base bit stream, (14; 40) and at least onesecond bit stream (15; 41, 42), subjecting the said at least one secondbit stream (15; 41, 42) to error control coding to produce at least oneoverlay bit stream (17; 44, 46), utilising said base and overlay bitstreams to generate a succession of symbol words of binary form, themost significant bit or bits of each symbol word being derived from thebase bit stream while the least significant bit or bits are derived fromthe said overlay bit stream, and utilising said symbol words to controlthe modulation level of a carrier signal during respective timeintervals, said modulation level during each said time intervalcomprising a base modulation level component defined by the mostsignificant bit or bits of the respective symbol word, and an overlaidmodulation level component defined by the least significant bit or bitsof said respective symbol word and linearly superposed upon said basemodulation level component.
 2. A method of data communication accordingto claim 1, further including receiving the transmitted data by areceiving process including the steps of:generating a succession ofmodulation-level binary words representing the level of modulation (φ)of the carrier signal during respective modulation time intervals,deriving from the least significant bits of the modulation-level wordsat least one received overlay bit stream (17; 44, 46) generated duringthe transmission process, effecting error correction on the said atleast one received overlay bit stream in accordance with the errorcontrol coding implemented in the transmission process, subtracting thebits of the said at least one overlay bit stream, after errorcorrection, from the modulation-level words and then utilising thelatter to derive a received base bit stream corresponding to the saidbase bit stream (14; 46) generated during the transmission process,utilising the said at least one error-corrected received overlay bitstream to generate at least one received second bit stream correspondingto said at least one second bit stream (15; 41, 42) generated during thetransmission process, and combining the received base bit stream and theat least one received second bit stream to form a received data stream.3. A method of data communication according to claim 1, wherein the basebit stream (14; 40) is also subject to error control coding before beingutilised, jointly with said at one overlay bit stream (17; 44, 46), togenerate said succession of symbol words.
 4. A method of datacommunication according to claim 1, wherein said at least one overlaybit stream (17; 44, 46) is divided into further bit streams at least oneof which is suject to further error control coding before being utilisedin the generation of said succession of symbol words.
 5. A method ofdata communication according to claim 1, wherein the error controlcoding to which the said at least one second bit stream (15; 41, 42) issubjected, utilises an orthogonal or bi-orthogonal block code.
 6. Amethod of data communication according to claim 5, wherein said binarydata stream is divided into said first, base, stream (40) and two saidsecond streams (41, 42), the second streams (41, 42) being error codedusing respective codes whereby to produce respective said overlay bitstreams (44, 46) which are then used one bit a time in generation ofsaid succession of symbol words, the codeword length of said codes usedin producing each overlay stream (44, 46) being the greater the lesserthe significance in each symbol word of the bit contributed by thatoverlay stream.
 7. A method of data communication according to claim 6,wherein the codeword length of the codes used in producing said overlaystreams (44, 46) changes by a factor of four between streams providingbits of adjacent significance in said symbol words.
 8. Datacommunication apparatus for transmitting a binary data stream,characterised in that said apparatus comprises:data-stream dividingmeans (8) for dividing the data stream into a first, base bit stream(14; 40) and at least one second bit stream (15; 41, 42), error-codingmeans (18; 43, 45) arranged to subject the said at least one second bitstream (15; 41, 42) to error control coding to produce at least oneoverlay bit stream (17; 44, 46), a signal organiser arranged to receivesaid base and overlay streams and to generate therefrom a succession ofsymbol words of binary form, the most significant bit or bits of eachsymbol word being derived from the base bit stream while the leastsignificant bit or bits are derived from said at least one overlay bitstream, and a modulator coupled to the signal organiser and arranged tomodulate a carrier signal such that the level of modulation iscontrolled according to said symbol words during respective modulationtime intervals, with said modulation level during each said timeinterval comprising a base modulation level component defined by themost significant bit or bits of the respective symbol word, and anoverlaid modulation component defined by the least significant bit orbits of said respective symbol word and linearly superposed upon saidbase modulation level component.
 9. Data communication apparatusarranged to receive data in the form of a modulated signal having amodulation level representative of a transmitted base bit stream and atleast one transmitted overlay bit stream which has been subjected toerror control coding, the said transmitted bit streams being derivedfrom an original data stream divided into two or more parts, wherein theapparatus comprises:demodulator means (20) for generating a successionof modulation level binary words representing the level of modulation ofthe carrier signal during respective modulation time intervals, overlaydetector means (21; 50, 21) for deriving from the least significant bitsof the said modulation-level words at least one received overlay bitstream corresponding to the said at least one overlay bit stream (17;44, 46) generated during the transmission process, the overlay detectormeans (21; 50, 21) being arranged to effect error correction on the saidat least one received overlay bit stream in accordance with the errorcontrol coding implemented in the transmission process, overlaysubtractor means (23; 52, 23) for removing the bits of the said at leastone overlay bit stream, after error correction, from themodulation-level words, base-stream decoder means (24) arranged toutilise said modulation-level words, after subtraction of said overlaystream bits, to derive a received base bit stream corresponding to thesaid base bit stream (14; 40) generated during the transmission process,overlay decoder means (25; 53, 25) arranged to utilise the said at leastone error-corrected received overlay bit stream to generate at least onereceived second bit stream corresponding to said at least one second bitstream (15; 41, 42) generated during the transmission process, andcombining means (19) for combining the received base bit stream and theat least one received second bit stream to form a received data stream.